DataSheet.jp

AN1042D の電気的特性と機能

AN1042DのメーカーはON Semiconductorです、この部品の機能は「High Fidelity Switching Audio Amplifiers Using TMOS Power MOSFETs」です。


製品の詳細 ( Datasheet PDF )

部品番号 AN1042D
部品説明 High Fidelity Switching Audio Amplifiers Using TMOS Power MOSFETs
メーカ ON Semiconductor
ロゴ ON Semiconductor ロゴ 




このページの下部にプレビューとAN1042Dダウンロード(pdfファイル)リンクがあります。

Total 12 pages

No Preview Available !

AN1042D Datasheet, AN1042D PDF,ピン配置, 機能
AN1042/D
High Fidelity Switching
Audio Amplifiers Using
TMOS Power MOSFETs
Prepared by: Donald E. Pauly
ON Semiconductor
Special Consultant
http://onsemi.com
APPLICATION NOTE
Almost all switching amplifiers operate by generating a
high frequency square wave of variable duty cycle. This
square wave can be generated much more efficiently than
an analog waveform. By varying the duty cycle from 0 to
100%, a net dc component is created that ranges between
the negative and positive supply voltages. A low pass filter
delivers this dc component to the speaker. The square wave
must be generated at a frequency well above the range of
hearing in order to be able to cover the full audio spectrum
from dc to 20 kHz. Figure 1 shows a square wave
generating a sine wave of one–ninth its frequency as its
duty cycle is varied.
1.0
0.75
0.5
0.25
0
–0.25
–0.5
–0.75
–1.0
Input
Output
Switching Frequency =
9X Modulation Frequency
+1
0
–1
0°
90°
180°
270°
360° 420°
Figure 1. Switching Amplifier Basic Waveforms
The concept of switching amplifiers has been around for
about 50 years but they were impractical before the advent
of complementary TMOS power MOSFETs. Vacuum tubes
were fast enough but they were rather poor switches. A
totem pole circuit with supply voltages of ±250 volts would
drop about 50 volts when switching a current of 200
milliamps. The efficiency of a tube switching amp could
therefore not exceed 80%. The transformer needed to
match the high plate impedance to the low impedance
speaker filter was impractical as well.
This document may contain references to devices which are no
longer offered. Please contact your ON Semiconductor represen-
tative for information on possible replacement devices.
With the introduction of complementary bipolar power
transistors in the late 1960s, switching amplifiers became
theoretically practical. At low frequencies, bipolar transistors
have switching efficiencies of 99% and will directly drive
a low impedance speaker filter. The requirement for
switching frequencies above 100 kHz resulted in excessive
losses however. Bipolar drive circuitry was also complex
because of its large base current requirement.
With the advent of complementary (voltage/current
ratings) TMOS power MOSFETs, gate drive circuitry has
been simplified. These MOS devices are very efficient as
switches and they can operate at higher frequencies.
A block diagram of the amplifier is shown in Figure 2.
An output switch connects either +44 or –44 volts to the
input of the low pass filter. This switch operates at a carrier
frequency of 120 kHz. Its duty cycle can vary from 5% to
95% which allows the speaker voltage to reach 90% of
either the positive or negative supplies. The filter has a
response in the audio frequency range that is as flat as
possible, with high attenuation of the carrier frequency and
its harmonics. A 0.05 ohm current sense resistor (R27) is
used in the ground return of the filter and speaker to provide
short circuit protection.
The negative feedback loop is closed before the filter to
prevent instabilities. Feedback cannot be taken from the
speaker because of the phase shift of the output filter, which
varies from 0° at dc to nearly 360° at 120 kHz. Since the
filter is linear, feedback may be taken from the filter input,
which has no phase shift. Unfortunately, this point is a high
frequency square wave which must be integrated to
determine its average voltage. The input is mixed with the
square wave output by resistors R4 and R5 shown in Figure
2. The resultant signal is integrated, which accurately
simulates the effect of the output filter. The output of the
integrator will be zero only if the filter input is an accurate
inverted reproduction of the amplifier input. If the output
is higher or lower than desired, the integrator will generate
a negative or positive error voltage. This error voltage is
applied to the input of the switch controller, which makes
the required correction. The integrator introduces a 90°
phase shift at high frequencies which leaves a phase margin
of nearly 90°.
© Semiconductor Components Industries, LLC, 2002
August, 2002 – Rev. 3
1
Publication Order Number:
AN1042/D
Free Datasheet http://www.datasheet4u.com/

1 Page





AN1042D pdf, ピン配列
AN1042/D
5
4
3
Parallel
2 Resonance
1 11.4 kHz
Parallel
0
Series
Resonance
–1
Resonance
35.2 kHz
20 kHz
–2
–3
–4
–5
0 12 24 36 48
kHz
Figure 4. Four Pole Butterworth Filter
Output Impedance
60
The amplifier output impedance at dc is about 4
milliohms and gradually becomes inductive. At 100 Hz, its
output impedance is 0.1 ohm giving a damping factor of 80.
Damping factor is the ratio of load impedance to amplifier
output impedance.
The complementary power MOSFET output stage of the
amplifier is shown in Figure 5. It generates a ±44 volt
square wave whose duty cycle can vary from 5% to 95%.
This variable duty cycle square wave is fed to the output
filter where the low frequency component is passed on to
the 8 ohm speaker. This filter allows frequencies under
20 kHz to pass with negligible loss, but greatly attenuates
the switching frequency. Since both sources are connected
to a supply rail, a drive of 10 volts peak to peak on each gate
insures full turn on. A buffer amp using ±5 volts supplies
provides this drive.
The 4.7 ohm resistors, R17 and R18, in each gate lead
prevents high frequency oscillation during switching. The 12
volt Zeners, CR3 and CR4, serve both as conventional
diode clamps and provide static discharge protection. They
act as dc restorers, and are made necessary by the ac
coupling. The 10 k resistors, R15 and R16, provide a slight
discharge path to keep conduction pulses in the clamp
diodes. They also discharge the gates in about 1
millisecond if the drive signal is lost. About 9 volts of
turn–on bias is applied to each gate. Tight coupling
between the gates prevents simultaneous turn–on of both
devices.
The output stage inverts the drive signal and generates
rise and fall times of about 30 nanoseconds. It is designed
to put out a maximum current of ±5 amps down to a
frequency of 0.1 Hertz. Below that frequency, maximum
current may need to be derated to prevent alternate
overheating of each output device. Excessive heatsink
temperature increases the ON resistance and the storage
time of the source drain diode. The resultant increase in
losses can lead to thermal runaway.
The drive waveform duty cycle must be a linear function
of the control voltage. The Duty Cycle Controller is shown
in Figure 6. A square wave of ±5 volts at 120 kHz is coupled
through C1 and R1 to integrator U1B. C1 blocks dc and R1
is the integrator resistor. C2 is the integrator capacitor
which generates a ±2 volt triangle on the output of U1B. R2
provides a small amount of dc leakage to insure that the
output has no significant dc component. R3 couples the
triangle to the noninverting input of comparator U1D. It
improves the waveform by isolating the input capacitance
of the comparator from the integrator. The dc offset on the
triangle is equal to the offset of U1B and its linearity is
better than 1%.
Input audio is applied to the inverting input of U2C
through R4. The output square wave of the power amp is
applied through R5 to the same summing point. U2C
functions as an integrator with C3 as the integrator
capacitor. Since R5 is 20 times R4, an inverting voltage
gain of 20 must result if the input of U2C is to be at ground.
The output of U2C serves as the error voltage and is fed to
the inverting input of U1D through R6 and R7. C4
eliminates short spikes on the error buss. Current limiting
circuitry is connected to the junction of R6 and R7. When
current drawn from the amplifier tries to exceed safe limits,
the error voltage is overridden and overcurrent is prevented.
+44
Drive
CR3 R15
C7 R17
C8
CR4
R18
R16
Q3
L1 L2
Q4
Feedback
C9 C10
R27
Current Sense
Figure 5. Output Circuit of a Class D Amplifier
http://onsemi.com
3
8
Speaker
Free Datasheet http://www.datasheet4u.com/


3Pages


AN1042D 電子部品, 半導体
AN1042/D
reduce the output to zero. At frequencies above 8 kHz, filter
phase shift makes the current limiting ineffective. This will
not be a problem unless the output is short circuited during
high frequency sine wave testing. R30 through R33 set the
bias currents for the operational amplifiers and
comparators.
The efficiency of a class B amplifier at the point of
clipping
is
p
4
or
78.5%.
As
the
output
is
reduced
the
efficiency linearly drops to 0% with no voltage out. The
average power of music integrated over one second has
been measured by the author at one tenth of peak power.
This does not seem to vary appreciably with different music
or speech as long as they are continuous. Under these
conditions, a class B amplifier will have an effective
efficiency of 25%. Figure 9 shows a plot of heatsink power
loss for a class B amplifier and a switching amplifier as a
function of output power. Note that a class B amplifier
actually runs hottest at slightly less than half power.
Maximum heating occurs at 40% of maximum power. The
heat rise varies only 25% as the power changes from 10%
to 90% of maximum.
As a result, a switching amplifier has one–tenth of the
heatsink requirements of a class B amplifier. Its greater
efficiency allows it to use a power supply of one–fourth the
size of a class B amplifier power supply. The author used
a switching power supply operated off 120 vac line at 20 kHz.
If a switching power supply is used, proper shielding must
be provided to prevent pickup of power supply spikes by
sensitive portions of the amplifier. A discussion of power
supplies is beyond the scope of this paper.
Switching amplifiers have a little known property of
power supply buss runaway when producing dc or low
frequency ac. The origin of this problem can be understood
by referring to Figure 10. It shows the current in the positive
switch when a sine wave just short of clipping is produced
by the amp. During the first half cycle, the switch is on most
of the time and power is delivered to the load, with some
energy being stored in the output inductor. During the
second half cycle, the switch is off most of the time and
current flow is reversed through the switch. This reverse
current comes from the output inductor, which is returning
energy to the positive supply through the source drain diode
of Q3. The forward current of the first half cycle tends to
drop the positive supply voltage, and the reverse current of
the second half cycle will raise it.
It can be shown that the current averaged during the
switching
cycle
in
the
positive
switch
is
sin
x
)
2
sin2Ăx.
This function is shown by the dashed curve in Figure 10.
The current averaged during the first half cycle of the
output
sine
wave
is
p
)
4p
4,
which
is
0.5683
times
peak
current. The average current during the second half output
of
the
cycle
is
p*
4p
4,
which
is
0.0683
times
peak.
0.25
Class B Amplifier
0.20
0.15
0.10
0.05 Theoretical Switching Amplifier
(0.3 Output Impedance, 8 Load)
0
0 0.2 0.4 0.6 0.8
Output Power (Normalized)
Figure 9. Power Loss versus Output Power
in Class B and Switching Amplifier
1.0
The average current during the complete cycle is 0.250,
being half the average power of a sine wave referenced to
peak power. Average reverse current through the switch peaks
at 0.125 of average peak forward current. This will cause
the voltages in a conventional supply to build to destructive
levels in short order, unless the power source is a battery.
Figure 10 only applies to a switching amp that is operated
just short of clipping with a normal load. If the amp is
operated into a short, conditions worsen. The average
forward and reverse currents will both be 0.500. This
means that no net power will be taken from the supply when
averaged over the whole cycle. This reflects the fact that no
power can be delivered into a short circuit.
To accommodate shorts on the output of the amplifier
without generating dangerous voltages, special power
supply circuitry must be used. These circuits must be able
to handle reverse currents on each buss equal to one–half
of the peak short circuit output current. Conventional
rectifier based supplies will not tolerate reverse current for
sustained periods. Large filter capacitors help but they
merely postpone the inevitable reckoning. A better
solution is coupling between the positive and negative
power supply busses.
1.0
0.75
0.5
0.25
0
–0.25
–0.5
–0.75
–1.0
sin x ) sin2 x
2
Positive Switch Current
Switching Frequency = 9X Modulation Frequency
Figure 10. Supply Buss Runaway
http://onsemi.com
6
Free Datasheet http://www.datasheet4u.com/

6 Page



ページ 合計 : 12 ページ
 
PDF
ダウンロード
[ AN1042D データシート.PDF ]


データシートを活用すると、その部品の主な機能と仕様を詳しく理解できます。 ピン構成、電気的特性、動作パラメータ、性能を確認してください。


共有リンク

Link :


部品番号部品説明メーカ
AN1042

High Fidelity Switching Audio Amplifiers Using TMOS Power MOSFETs

ON Semiconductor
ON Semiconductor
AN1042D

High Fidelity Switching Audio Amplifiers Using TMOS Power MOSFETs

ON Semiconductor
ON Semiconductor


www.DataSheet.jp    |   2020   |  メール    |   最新    |   Sitemap